This article is devoted to the 2161 Second Edition (SE) series of switching power supplies based on the IR2161 controller.
- Short circuit and overload protection;
- Auto reset short circuit protection;
- Frequency modulation "dither" (to reduce EMI);
- Microcurrent startup (for initial startup of the controller, a current of no more than 300 μA is sufficient);
- Possibility of dimming (but we are not interested in this);
- Output voltage compensation (a kind of voltage stabilization);
- Soft start;
- Adaptive dead time ADT;
- Compact body;
- Produced using lead-free technology (Leed-Free).
I will give some important ones for us specifications:
Maximum inflow/outflow current: +/-500mA
A sufficiently large current allows you to control powerful switches and build quite powerful switching power supplies based on this controller without the use of additional drivers;
Maximum current consumed by the controller: 10mA
Based on this value, the power circuits of the microcircuit are designed;
Minimum operating voltage of the controller: 10.5V
At a lower supply voltage, the controller switches to UVLO mode and the oscillation stops;
Minimum stabilization voltage of the zener diode built into the controller: 14.5V
The external zener diode must have a stabilization voltage no higher than this value to avoid damage to the microcircuit due to shunting excess current to the COM pin;
Voltage at the CS pin to trigger overload protection: 0.5V
The minimum voltage at the CS pin at which the overload protection is triggered;
Voltage at the CS pin for short circuit protection: 1V
The minimum voltage at the CS pin at which short circuit protection is triggered;
Operating frequency range: 34 - 70 kHz
The operating frequency is not directly set and depends only on the power consumed by the load;
Default dead time: 1µS
Used when it is impossible to work in adaptive dead time (ADT) mode, as well as when there is no load;
Operating frequency in soft start mode: 130 kHz
The frequency at which the controller operates in soft start mode;
The main attention should now be paid to what operating modes of the microcircuit exist and in what sequence they are located relative to each other. I will focus on describing the operating principle of each of the circuit blocks, and I will describe the sequence of their operation and the conditions for transition from one mode to another more briefly. I'll start with a description of each of the blocks of the diagram:
Under-voltage Lock-Out Mode (UVLO)- the mode in which the controller is when its supply voltage is below the minimum threshold value (approximately 10.5V).
Soft Start Mode- operating mode in which the controller oscillator operates at an increased frequency for a short time. When the oscillator is turned on, its operating frequency is initially very high (about 130 kHz). This causes the converter output voltage to be lower because the power supply transformer has a fixed inductance which will have a higher impedance at higher frequency and thus reduces the voltage on the primary winding. Reduced voltage will naturally result in reduced current in the load. As the CSD capacitor charges from 0 to 5V, the oscillation frequency will gradually decrease from 130 kHz to the operating frequency. The duration of the soft start sweep will depend on the capacitance of the CSD capacitor. However, since the CSD capacitor also sets the shutdown delay time and participates in the operation of the voltage compensation unit, its capacitance must be strictly 100nF.
Soft start problem. I would like to be completely honest and mention the fact that if there are high-capacity filter capacitors at the output of the power supply, soft start most often does not work and the SMPS starts immediately at the operating frequency, bypassing the soft start mode. This happens due to the fact that at the moment of start, the discharged capacitors in the secondary circuit have a very low intrinsic resistance and a very high current is required to charge them. This current causes a short-circuit protection to operate briefly, after which the controller immediately restarts and goes into RUN mode, bypassing the soft start mode. You can combat this by increasing the inductance of the chokes in the secondary circuit, located immediately after the rectifier. Chokes with high inductance extend the charging process of the output filter capacitors; in other words, the capacitors are charged with a smaller current, but longer in time. A lower charging current does not trigger the protection at start and allows the soft start to perform its functions normally. Just in case, regarding this issue, I contacted the manufacturer’s technical support, to which I received the following answer:
"A typical halogen converter has an AC output without rectifiers or output capacitors. Soft starting works by reducing the frequency. To achieve soft starting, the transformer needs to have significant leakage. However, this should be possible in your case. Try placing the inductor on the secondary side of the bridges diodes to the capacitor.
Best wishes.
Infineon Technologies
Steve Rhyme, Support Engineer"
My assumptions about the reason for the unstable operation of soft start turned out to be correct and, moreover, they even offered me the same method of dealing with this problem. And again, to be completely honest, it should be added that the use of coils with increased inductance, relative to those usually used at the output of the SMPS, improves the situation, but does not completely eliminate the problem. However, this problem can be tolerated given that there is a thermistor at the SMPS input that limits the inrush current.
Run Mode, operating mode. When the soft start is completed, the system enters voltage compensated operating mode. This function provides some stabilization of the converter output voltage. Voltage compensation occurs by changing the operating frequency of the converter (increasing the frequency reduces the output voltage), although the accuracy of this type of “stabilization” is not high, it is nonlinear and depends on many parameters and, therefore, is not easy to predict. IR2161 monitors the load current through a current resistor (RCS). The peak current is detected and amplified in the controller and then applied to the CSD pin. The voltage on the CSD capacitor, in operating mode (voltage compensation mode), will vary from 0 (at minimum load) to 5V (at maximum load). In this case, the generator frequency will vary from 34 kHz (Vcsd = 5V) to 70 kHz (Vcsd = 0V).
It is also possible to attach feedback to the IR2161, which will allow you to organize almost complete stabilization of the output voltage and will allow you to much more accurately monitor and maintain the required voltage at the output: 
We will not consider this scheme in detail within the framework of this article.
Shut Down Mode, shutdown mode. The IR2161 contains a two-position automatic shutdown system that detects both short circuit and overload conditions of the inverter. The voltage at the CS pin is used to determine these conditions. If the output of the converter is shorted, a very large current will flow through the switches and the system must shut down within a few periods of time on the grid, otherwise the transistors will be quickly destroyed due to thermal runaway of the junction. The CS pin has a turn-off delay to prevent nuisance tripping, either due to inrush current at turn-on or due to transient currents. Lower threshold (when Vcs > 0.5< 1 В), имеет намного большую задержку до отключения ИИП. Задержка для отключения по перегрузке приблизительно равна 0,5 сек. Оба режима отключения (по перегрузке и по короткому замыканию), имеют автоматический сброс, что позволяет контроллеру возобновить работу примерно через 1 сек после устранения перегрузки или короткого замыкания. Это значит, что если неисправность будет устранена, преобразователь может продолжить нормально работать. Осциллятор работает на минимальной рабочей частоте (34 кГц), когда конденсатор CSD переключается к цепи отключения. В режиме плавного пуска или рабочем режиме, если превышен порог перегрузки (Vcs >0.5V), IR2161 quickly charges CSD up to 5V. When the voltage at the CS pin is greater than 0.5V and when the short circuit threshold of 1V is exceeded, the CSD will charge from 5V to the controller supply voltage (10-15V) in 50ms. When the overload threshold voltage Vcs is more than 0.5V but less than 1V, the CSD is charged from 5V to the supply voltage in approximately 0.5 sec. It should be remembered and taken into account the fact that high-frequency pulses with a 50% duty cycle and a sinusoidal envelope appear at the CS pin - this means that only at the peak of the network voltage the CSD capacitor will be charged in stages, in each half-cycle. When the voltage on the CSD capacitor reaches the supply voltage, the CSD is discharged to 2.4V and the converter starts again. If the fault is still present, the CSD starts charging again. If the fault disappears, the CSD will discharge to 2.4V, and then the system will automatically return to the voltage compensation operating mode.
STANDBY mode, standby mode- the mode in which the controller is in the case of insufficient supply voltage, while it consumes no more than 300 μA. In this case, the oscillator is naturally turned off and the SMPS does not work; there is no voltage at its output.
Blocks Fault Timing Mode, Delay and Fault Mode, although shown in the block diagram, are not essentially operating modes of the controller; rather, they can be attributed to transition stages (Delay and Fault Mode) or conditions for transition from one mode to another (Fault Timing Mode).
Now I’ll describe how does it all work together:
When power is applied, the controller starts in UVLO mode. As soon as the controller supply voltage exceeds the minimum voltage value required for stable operation, the controller switches to soft start mode, the oscillator starts at a frequency of 130 kHz. The CSD capacitor charges smoothly up to 5V. As the external capacitors charge, the operating frequency of the oscillator decreases to the operating frequency. Thus, the controller switches to RUN mode. As soon as the controller enters RUN mode, the CSD capacitor is instantly discharged to ground potential and is connected by an internal switch to the voltage compensation circuit. If the SMPS is started not at idle, but under load, there will be a potential at the CS pin proportional to the load value, which, through the internal circuits of the controller, will affect the voltage compensation unit and will not allow the CSD capacitor, after the completion of the soft start, to completely discharge. Thanks to this, the start will not occur at the maximum frequency of the operating range, but at a frequency corresponding to the load value at the output of the SMPS. After switching to RUN mode, the controller works according to the situation: either it remains working in this mode until you get tired and unplug the power supply from the outlet, or... In case of overheating, the controller goes into FAULT mode, the oscillator stops working . After the chip cools down, a restart occurs. In the event of an overload or short circuit, the controller goes into Fault Timing mode, and the external capacitor CSD is instantly disconnected from the voltage compensation unit and connected to the shutdown unit (the CSD capacitor in this case sets the controller shutdown delay time). The operating frequency is instantly reduced to the minimum. In case of overload (when the voltage at the CS pin > 0.5< 1 В), контроллер переходит в режим SHUTDOWN и выключается, но происходит это не мгновенно, а только в том случае, если перегрузка продолжается дольше половины секунды. Если перегрузки носят импульсный характер с продолжительностью импульса не более 0,5 сек, то контроллер будет просто работать на минимально возможно частоте, постоянно переключаясь между режимами RUN, Fault Timing, Delay, RUN (при этом будут отчетливо слышны щелчки). Когда напряжение на выводе CS превышает 1В, срабатывает защита от короткого замыкания. При устранении перегрузки или короткого замыкания, контроллер переходит в режим STANDBY и при наличии благоприятных условий для перезапуска, минуя режим софт-старта, переходит в режим RUN.
Now that you understand how the IR2161 works (I hope so), I will tell you about the switching power supplies themselves based on it. I want to immediately warn you that if you decide to assemble a switching power supply based on this controller, then you should assemble the SMPS guided by the latest, most advanced circuit on the corresponding printed circuit board. Therefore, the list of radio elements at the bottom of the article will be given only for the latest version of the power supply. All intermediate editions of the IIP are shown only to demonstrate the process of improving the device.
And the first IIP that will be discussed is conventionally named by me 2161 SE 2.
The main and key difference of the 2161 SE 2 is the presence of a controller self-supply circuit, which made it possible to get rid of boiling quenching resistors and, accordingly, increase the efficiency by several percent. Other equally significant improvements were also made: optimization of the printed circuit board layout, more output terminals were added for connecting the load, and a varistor was added.
The SMPS diagram is shown in the image below:
The self-powering circuit is built on VD1, VD2, VD3 and C8. Due to the fact that the self-supply circuit is connected not to a low-frequency 220V network (with a frequency of 50Hz), but to the primary winding of a high-frequency transformer, the capacity of the self-supply quenching capacitor (C8) is only 330pF. If self-supply was organized from a low-frequency 50Hz network, then the capacity of the quenching capacitor would have to be increased 1000 times, and of course such a capacitor would take up much more space on the printed circuit board. The described method of self-powering is no less effective than self-powering from a separate winding of a transformer, but it is much simpler. Zener diode VD1 is necessary to facilitate the operation of the built-in zener diode of the controller, which is not capable of dissipating significant power and without installing an external zener diode can simply be broken, which will lead to a complete loss of functionality of the microcircuit. The stabilization voltage VD1 should be in the range of 12 - 14V and should not exceed the stabilization voltage of the controller's built-in zener diode, which is approximately 14.5V. As VD1, you can use a zener diode with a stabilization voltage of 13V (for example, 1N4743 or BZX55-C13), or use several zener diodes connected in series, which is what I did. I connected two zener diodes in series: one of them was 8.2V, the other was 5.1V, which ultimately gave a resulting voltage of 13.3V. With this approach to powering the IR2161, the controller’s supply voltage does not sags and is practically independent of the load size connected to the SMPS output. In this scheme, R1 is only needed to start the controller, so to speak, for the initial kick. R1 gets a little warm, but not nearly as much as it was in the first version of this power supply. The use of high-resistance resistor R1 provides another interesting feature: the voltage at the output of the SMPS does not appear immediately after being connected to the network, but after 1-2 seconds, when C3 is charged to the minimum voltage of 2161 (approximately 10.5V).
Starting with this SMPS and all subsequent ones, a varistor is used at the SMPS input; it is designed to protect the SMPS from exceeding the input voltage above the permissible value (in this case - 275V), and also very effectively suppresses high-voltage interference by preventing them from entering the SMPS input from network and without releasing interference from the SMPS back into the network.
In the rectifier of the secondary power supply of the power supply, I used SF54 diodes (200V, 5A) two in parallel. The diodes are located on two floors, the leads of the diodes should be as long as possible - this is necessary for better heat dissipation (the leads are a kind of radiator for the diode) and better air circulation around the diodes.
The transformer in my case is made on a core from a computer power supply - ER35/21/11. The primary winding has 46 turns in three 0.5mm wires, two secondary windings have 12 turns in three 0.5mm wires. The input and output chokes are also taken from the computer power supply.
The described power supply is capable of delivering 250W to the load for a long time (without operating time limitation), and 350W for a short time (no more than a minute). When using this SMPS in dynamic load mode (for example, to power an audio frequency power amplifier of class B or AB), it is possible to power an UMZCH with a total output power of 300W (2x150W in stereo mode) from this switching power supply.
Oscillogram on the primary winding of the transformer (without snubber, R5 = 0.15 Ohm, 190W output):
As can be seen from the oscillogram, with an output power of 190 W, the operating frequency of the SMPS is reduced to 38 kHz; at idle, the SMPS operates at a frequency of 78 kHz:
From the oscillograms, in addition, it is clearly visible that there are no outliers on the graph, and this undoubtedly characterizes this SMPS positively.
At the output of the power supply, in one of the arms you can see the following picture:
The ripple has a frequency of 100Hz and a ripple voltage of approximately 0.7V, which is comparable to the ripple at the output of a classic, linear, non-stabilized power supply. For comparison, here is an oscillogram taken when operating at the same output power for a classic power supply (capacitor capacity 15000 μF in the arm):
As can be seen from the oscillograms, the supply voltage ripple at the output of a switching power supply is lower than that of a classic power supply of the same power (0.7V for an SMPS, versus 1V for a classic unit). But unlike a classic power supply, a small high-frequency noise is noticeable at the output of the SMPS. However, there is no significant high-frequency interference or emissions. The ripple frequency of the supply voltage at the output is 100Hz and is caused by the voltage ripple in the primary circuit of the SMPS along the +310V bus. To further reduce ripple at the SMPS output, it is necessary to increase the capacitance of capacitor C9 in the primary circuit of the power supply or the capacitance of the capacitors in the secondary circuit of the power supply (the former is more effective), and to reduce high-frequency interference, use chokes with higher inductance at the SMPS output.
The PCB looks like this:
The following SMPS diagram that will be discussed is 2161 SE 3:
The finished power supply assembled according to this diagram looks like this:
There are no fundamental differences in the circuit from SE 2; the differences mainly concern the printed circuit board. The circuit added only snubbers in the secondary windings of the transformer - R7, C22 and R8, C23. The values of the gate resistors have been increased from 22 Ohm to 51 Ohm. The value of capacitor C4 has been reduced from 220 µF to 47 µF. Resistor R1 is assembled from four 0.5W resistors, which made it possible to reduce the heating of this resistor and make the design slightly cheaper because In my area, four half-watt resistors are cheaper than one two-watt one. But the opportunity to install one two-watt resistor remains. In addition, the value of the self-feeding capacitor was increased to 470pF, there was no particular point in this, but it was done as an experiment, the flight was normal. MUR1560 diodes in a TO-220 package are used as rectifier diodes in the secondary circuit. Optimized and reduced printed circuit board. The dimensions of the SE 2 printed circuit board are 153x88, while the SE 3 printed circuit board has dimensions of 134x88. The PCB looks like this:
The transformer is made on a core from a computer power supply - ER35/21/11. The primary winding has 45 turns in three 0.5mm wires, two secondary windings have 12 turns in four 0.5mm wires. The input and output chokes are also taken from the computer power supply.
The very first inclusion of this SMPS in the network showed that the snubbers in the secondary circuit of the power supply were clearly superfluous; they were immediately soldered off and were not used further. Later the snubber of the primary winding was also soldered off, as it turned out it did much more harm than good.
It was possible to extract 300-350W of power from this power supply for a long time; for a short time (no more than a minute) this SMPS can supply up to 500W; after a minute of operation in this mode, the overall radiator heats up to 60 degrees.
Look at the oscillograms:
Everything is still beautiful, the rectangle is almost perfectly rectangular, there are no outliers. With snubbers, oddly enough, everything was not so beautiful.
The following diagram is the final and most advanced 2161 SE 4:
When assembled, the device according to this diagram looks like this:
Like last time, there were no major changes in the scheme. Perhaps the most noticeable difference is that the snubbers have disappeared, both in the primary circuit and in the secondary ones. Because, as my experiments have shown, due to the peculiarities of the IR2161 controller, snubbers only interfere with its operation and are simply contraindicated. Other changes were also made. The values of the gate resistors (R3 and R4) have been reduced from 51 to 33 Ohms. In series with the self-feeding capacitor C7, a resistor R2 is added to protect against overcurrents when charging capacitors C3 and C4. Resistor R1 still consists of four half-watt resistors, and resistor R6 is now hidden under the board and consists of three SMD resistors of the 2512 format. Three resistors provide the required resistance, but it is not necessary to use exactly three resistors; depending on the required power, you can use one, two or three resistors are acceptable. Thermistor RT1 has been moved from the SMPS to the +310V target. The remaining measurements concern only the layout of the printed circuit board and it looks like this:
A safety gap has been added to the printed circuit board between the primary and secondary circuits, and a through cut has been made in the board at the narrowest point.
The transformer is exactly the same as in the previous power supply: it is made on a core from a computer power supply - ER35/21/11. The primary winding has 45 turns in three 0.5mm wires, two secondary windings have 12 turns in four 0.5mm wires. The input and output chokes are also taken from the computer power supply.
The output power of the power supply remained the same - 300-350W in long-term mode and 500W in short-term mode (no more than a minute). From this SMPS you can power a UMZCH with a total output power of up to 400W (2x200W in stereo mode).
Now let's look at the oscillograms on the primary winding of the transformer of this switching power supply:
Everything is still beautiful: the rectangle is rectangular, there are no outliers.
At the output of one of the arms of the power supply, at idle, you can observe the following picture:
As you can see, the output contains negligible high-frequency noise with a voltage of no more than 8 mV (0.008 V).
Under load, at the output, you can observe the already well-known ripples with a frequency of 100 Hz:
With an output power of 250W, the ripple voltage at the output of the SMPS is 1.2V, which, considering the lower capacitance of the capacitors in the secondary circuit (2000uF in the shoulder, versus 3200uF for SE2) and the high output power at which the measurements were made, looks very good. The high-frequency component at a given output power (250W) is also insignificant, has a more ordered character and does not exceed 0.2V, which is a good result.
Setting the protection threshold. The threshold at which the protection will operate is set by resistor RCS (R5 - in SE 2, R6 - in SE 3 and SE 4).
This resistor can be either output or SMD format 2512. RCS can be composed of several resistors connected in parallel.
The RCS denomination is calculated using the formula: Rcs = 32 / Pnom. Where, Pnom is the output power of the SMPS, above which the overload protection will operate.
Example: let's say that we need the overload protection to be triggered when the output power exceeds 275W. We calculate the resistor value: Rcs=32/275=0.116 Ohm. You can use either one 0.1 Ohm resistor, or two 0.22 Ohm resistors connected in parallel (which will result in 0.11 Ohm), or three 0.33 Ohm resistors, also connected in parallel (which will result in 0.11 Ohm) .
Now it’s time to touch on the topic that interests people the most - calculation of a transformer for a switching power supply. Due to your numerous requests, I will finally tell you in detail how to do this.
First of all, we need a core with a frame, or just a core if it is a ring-shaped core (shape R).
Cores and frames can be of completely different configurations and can be used in any way. I used an ER35 frame core from a computer power supply. The most important thing is that the core does not have a gap; cores with a gap cannot be used.
By default, immediately after starting the program, you will see similar numbers.
Starting the calculation, the first thing we will do is select the shape and dimensions of the core in the upper right corner of the program window. In my case, the shape is ER, and the sizes are 35/21/11.
The dimensions of the core can be measured independently; how to do this can be easily understood from the following illustration:
Next, select the core material. It’s good if you know what material your core is made of, if not, then it’s okay, just choose the default option - N87 Epcos. In our conditions, the choice of material will not have a significant impact on the final result.
The next step is to select the converter circuit; ours is half-bridge:
In the next part of the program - “supply voltage”, select “variable” and indicate 230V in all three windows.
In the “converter characteristics” part, we indicate the bipolar output voltage we need (voltage of one arm) and the required output power of the SMPS, as well as the diameter of the wire with which you want to wind the secondary and primary windings. In addition, the type of rectifier used is selected - “bipolar with a midpoint”. There we also check the box “use the desired diameters” and under “stabilization of outputs” select “no”. Select the type of cooling: active with a fan or passive without it. You should end up with something like this:
The actual values of the output voltages will be greater than what you indicate in the program when calculating. In this case, with a voltage of 2x45V specified in the program, the output of a real SMPS will be approximately 2x52V, so when calculating, I recommend specifying a voltage that is 3-5V less than required. Or indicate the required output voltage, but wind one turn less than indicated in the program calculation results. The output power should not exceed 350W (for 2161 SE 4). The diameter of the wire for winding, you can use any one you have, you need to measure and indicate its diameter. You should not wind the windings with a wire with a diameter of more than 0.8 mm; it is better to wind the windings using several (two, three or more) thin wires than one thick wire.
After all this, click on the “calculate” button and get the result, in my case it turned out like this:
We focus our attention on the points highlighted in red. The primary winding in my case will consist of 41 turns, wound in two wires with a diameter of 0.5 mm each. The secondary winding consists of two halves of 14 turns, wound in three wires with a diameter of 0.5 mm each.
After receiving all the necessary calculation data, we proceed directly to winding the transformer.
It seems to me that there is nothing complicated here. I'll tell you how I do it. First, the entire primary winding is wound. One of the ends of the wire(s) is stripped and soldered to the corresponding terminal of the transformer frame. After which the winding begins. The first layer is wound and then a thin layer of insulation is applied. After which the second layer is wound and a thin layer of insulation is applied again and thus the entire required number of turns of the primary winding is wound. It is best to wind the windings turn to turn, but you can also do it askew or just “anyhow”, this will not play a noticeable role. After the required number of turns have been wound, the end of the wire(s) is cut off, the end of the wire is stripped and soldered to another corresponding terminal of the transformer. After winding the primary winding, a thick layer of insulation is applied to it. It is best to use a special Mylar tape as insulation:
The same tape is used to insulate the windings of pulse transformers of computer power supplies. This tape conducts heat well and has high heat resistance. From available materials, it is recommended to use: FUM tape, masking tape, paper plaster or a baking sleeve cut into long strips. It is strictly forbidden to use PVC and fabric insulating tape, stationery tape, or fabric plaster to insulate windings.
After the primary winding is wound and insulated, we proceed to winding the secondary winding. Some people wind two halves of the winding at once and then separate them, but I wind the halves of the secondary winding one by one. The secondary winding is wound in the same way as the primary. First, we strip and solder one end of the wire(s) to the corresponding terminal of the transformer frame, wind the required number of turns, applying insulation after each layer. Having wound the required number of turns of one half of the secondary winding, we strip and solder the end of the wire to the corresponding terminal of the frame and apply a thin layer of insulation. We solder the beginning of the wire of the next half of the winding to the same terminal as the end of the previous half of the winding. We wind in the same direction, the same number of turns as the previous half of the winding, applying insulation after each layer. Having wound the required number of turns, solder the end of the wire to the corresponding terminal of the frame and apply a thin layer of insulation. There is no need to apply a thick layer of insulation after winding the secondary winding. At this point, the winding can be considered complete.
After winding is completed, it is necessary to insert the core into the frame and glue the core halves together. For gluing, I use one-second super glue. The adhesive layer should be minimal so as not to create a gap between the parts of the core. If you have a ring core (shape R), then naturally you won’t have to glue anything, but the winding process will be less convenient and will take more effort and nerves. In addition, the ring core is less convenient due to the fact that you will have to create and mold the transformer leads yourself, as well as think about attaching the finished transformer to the printed circuit board.
Upon completion of winding and assembly of the transformer, you should get something like this:
For convenience of narration, I will also add here the SMPS 2161 SE 4 diagram for a brief description talk about the element base and possible replacements.
Let's go in order - from entrance to exit. At the input, the mains voltage meets fuse F1; the fuse can have a rating from 3.15A to 5A. Varistor RV1 must be designed for 275V, such a varistor will be marked 07K431, but it is also possible to use variators 10K431 or 14K431. It is also possible to use a varistor with a higher threshold voltage, but the effectiveness of protection and noise suppression will be noticeably lower. Capacitors C1 and C2 can be either regular film capacitors (such as CL-21 or CBB-21) or noise-suppressing type (for example X2) for a voltage of 275V. We unsolder the dual inductor L1 from a computer power supply or other faulty equipment. The inductor can be made independently by winding 20-30 turns on a small ring core, with a wire with a diameter of 0.5 - 0.8 mm. The VDS1 diode bridge can be any for a current from 6 to 8A, for example, indicated in the diagram - KBU08 (8A) or RS607 (6A). Any slow or fast diode with a current from 0.1 to 1A and a reverse voltage of at least 400V is suitable as VD4. R1 can consist of either four half-watt resistors of 82 kOhm, or be one two-watt resistor with the same resistance. Zener diode VD1 must have a stabilization voltage in the range of 13 - 14V; it is allowed to use either one zener diode or a series connection of two zener diodes with a lower voltage. C3 and C5 can be either film or ceramic. C4 should have a capacitance of no more than 47 µF, voltage 16-25V. Diodes VD2, VD3, VD5 must be very fast, for example - HER108 or SF18. C6 can be either film or ceramic. Capacitor C7 must be designed for a voltage of at least 1000V. C9 can be either film or ceramic. The R6 rating must be calculated for the required output power, as described above. As R6, you can use either SMD resistors of the 2512 format or output one- or two-watt resistors; in any case, the resistor(s) are installed under the board. Capacitor C8 must be film (type CL-21 or CBB-21) and have an allowable operating voltage of at least 400V. C10 is an electrolytic capacitor with a voltage of at least 400V; the magnitude of low-frequency ripples at the output of the SMPS depends on its capacitance. RT1 is a thermistor, you can buy it, or you can unsolder it from a computer power supply, its resistance should be from 10 to 20 Ohms and the permissible current should be at least 3A. Both the IRF740 indicated in the diagram and other transistors with similar parameters, for example, IRF840, 2SK3568, STP10NK60, STP8NK80, 8N60, 10N60, can be used as transistors VT1 and VT2. Capacitors C11 and C13 must be film (type CL-21 or CBB-21) with a permissible voltage of at least 400V, their capacitance must not exceed the 0.47 μF indicated in the diagram. C12 and C14 are ceramic, high-voltage capacitors for a voltage of at least 1000V. The VDS2 diode bridge consists of four diodes connected by a bridge. As VDS2 diodes, it is necessary to use very fast and powerful diodes, for example, such as - MUR1520 (15A, 200V), MUR1560 (15A, 600V), MUR820 (8A, 200V), MUR860 (8A, 600V), BYW29 (8A, 200V) , 8ETH06 (8A, 600V), 15ETH06 (15A, 600V). Chokes L2 and L3 are soldered from the computer power supply or made independently. They can be wound either on individual ferrite rods or on a common ring core. Each of the chokes should contain from 5 to 30 turns (more is better), with a wire with a diameter of 1 - 1.5 mm. Capacitors C15, C17, C18, C20 must be film (type CL-21 or CBB-21) with a permissible voltage of 63V or more, the capacitance can be any, the larger their capacitance, the better, the stronger the suppression of high-frequency interference. Each of the capacitors designated in the diagram as C16 and C19 consists of two 1000uF 50V electrolytic capacitors. In your case you may need to use higher voltage capacitors.
And as a final touch, I’ll show you a photo that shows the evolution of the switching power supplies I created. Each subsequent SMPS is smaller, more powerful and better quality than the previous one: 
That's all! Thank you for your attention!
List of radioelements
| Designation | Type | Denomination | Quantity | Note | Shop | My notepad | |
|---|---|---|---|---|---|---|---|
| Switching Power Supply 2161 SE 4 | |||||||
| R1 | Resistor | 82 kOhm | 4 | 0.5W | To notepad | ||
| R2 | Resistor | 4.7 Ohm | 1 | 0.25W | To notepad | ||
| R3, R4 | Resistor | 33 Ohm | 2 | 0.25W | To notepad | ||
| R5 | Resistor | ||||||
Nowadays, it is rare that anyone introduces a mains transformer into a home-made amplifier design, and rightly so - a switching power supply is cheaper, lighter and more compact, and a well-assembled one produces almost no interference into the load (or interference is minimized).
Of course, I don’t argue that a network transformer is much, much more reliable, although modern impulse generators, stuffed with all kinds of protections, also do a good job of their task.
IR2153 is, I would say, a legendary microcircuit that is used very often by radio amateurs and is being implemented specifically in network switching power supplies. The microcircuit is a simple half-bridge driver and in power supply circuits it works as a pulse generator.
Based on this microcircuit, power supplies from several tens to several hundred watts and even up to 1500 watts are built; of course, as the power increases, the circuit will become more complicated.
Nevertheless, I don’t see the point in making a high-power power supply using this particular microcircuit, the reason is that it is impossible to organize output stabilization or control, and not only the microcircuit is not a PWM controller, therefore there can be no talk of any PWM control, and this is very bad . Good power supplies are usually made on push-pull PWM microcircuits, for example TL494 or its relatives, etc., and the block on IR2153 is more of a beginner-level block.
Let's move on to the design of the switching power supply itself. Everything is assembled according to the datasheet - a typical half-bridge, two half-bridge capacitors, which are constantly in a charge/discharge cycle. The power of the circuit as a whole will depend on the capacity of these capacitors (well, of course, not only on them). The calculated power of this particular option is 300 watts, I don’t need more, the unit itself is for powering two UHF channels. The capacity of each capacitor is 330 μF, the voltage is 200 Volts, any computer power supply contains just such capacitors, in theory, the circuit diagram of computer power supplies and our unit is somewhat similar, in both cases the topology is half-bridge.
At the input of the power supply, everything is also as it should be - a varistor for surge protection, a fuse, a surge protector and, of course, a rectifier. A full-fledged diode bridge, which you can take ready-made, the main thing is that the bridge or diodes have a reverse voltage of at least 400 Volts, ideally 1000, and with a current of at least 3 Amperes. Separating capacitor - film, 250 V or better 400, capacity 1 μF, by the way - can also be found in a computer power supply.
Transformer Calculated according to the program, the core is from a computer power supply unit, alas, I cannot indicate the overall dimensions. In my case, the primary winding is 37 turns with a 0.8mm wire, the secondary winding is 2 x 11 turns with a bus of 4 0.8mm wires. With this situation, the output voltage is around 30-35 Volts, of course, the winding data will be different for everyone, depending on the type and overall dimensions of the core.
I present to your attention the circuit I tested of a fairly simple switching power supply unit UMZCH. The power of the unit is about 200W (but can be overclocked to 500W).
Brief characteristics:
Input voltage - 220V;
Output voltage - +-26V (drawdown 2-4V at full load);
Conversion frequency - 100 kHz;
The maximum load current is 4A.
Block diagram
The power supply is built on the IR2153 chip according to the strannicmd circuit
Construction and details.
The power supply is assembled on a printed circuit board made of single-sided fiberglass. You will find a drawing of a printed circuit board in Sprint-Layout for an iron at the end of the article.
An input inductor from any computer or monitor power supply, an input capacitor is used at the rate of 1 µF per 1 W. Next, a flat low-frequency diode bridge GBUB of approximately 3A can be used as switches IRF 840, IRFI840GLC, IRFIBC30G, VT1 - BUT11, VT3 - c945, output diodes it is better to use assemblies more quickly in this circuit, I installed Schottky MBR 1545, the output chokes are made of pieces of ferrite 4 cm and ? 3 mm long, 26 turns of PEV-1 wire, but I think that you can also use a group stabilization choke on a ring of atomized iron (the haven't tried it).
Most of the parts can be found in computer power supplies.
Printed circuit board
PSU assembly
Transformer
Transformer for your needs, you can calculate
This transformer is wound on one K32X19X16 ring made of M2000NM ferrite (blue ring), the primary winding is wound evenly across the entire ring and is 34 turns of MGTF 0.7 wire. Before winding the secondary windings, you need to wrap the primary winding with fluoroplastic tape. Winding II is evenly wound with PEV-1 0.7 wire folded in half and is 6+6 turns with a tap from the middle. Winding III (self-powered IR) is uniformly wound 3+3 turns with twisted pair (one pair of wires) with a tap from the middle.
Setting up power supply
ATTENTION!!! THE PRIMARY CIRCUIT OF THE PSU IS UNDER MAINS VOLTAGE, SO PRECAUTIONS SHOULD BE FOLLOWED WHEN SETTING UP AND OPERATING.
It is advisable to start the unit for the first time by connecting it through a current-limiting resistor to the fuse, which is an incandescent lamp with a power of 60 W and a voltage of 220 V, and the IR should be powered from a separate 12 V power supply (the self-supply winding is turned off). When the power supply is turned on, do not load it heavily through the lamp. As a rule, a correctly assembled power supply does not require adjustment. When you turn it on for the first time through the power supply lamp, the lamp should light up and immediately go out (blink), but if so, then everything is fine and you can check the power at the output. All OK! then we turn off the lamp, set the fuse and connect the self-power of the microcircuit; when the power supply starts, the LED that is located between the first and third legs should blink and the power supply will start.
Hello everyone!!!
I present to your attention the circuit I tested of a fairly simple switching power supply unit UMZCH. The power of the unit is about 180 W.
Brief characteristics of the UPS
Input voltage - 220V;Output voltage - ±25V;
Conversion frequency - 27 kHz;
Maximum load current - 3.5A.
Switching power supply circuit
The scheme is quite simple:It is a half-bridge inverter with a switching saturable transformer. Capacitors C1 and C2 form a voltage divider for one half of the half-bridge, and also smooth out the ripples of the mains voltage. The second half of the half-bridge is transistors VT1 and VT2, controlled by switching transformer T2. The diagonal of the bridge includes the primary winding of the power transformer T1, which is designed so that it does not become saturated during operation.
To reliably start the converter, a relaxation generator is used on a VT3 transistor operating in avalanche mode.
Briefly the principle of its operation. Capacitor C7 is charged through resistor R3, while the voltage at the collector of transistor VT3 increases in a sawtooth manner. When this voltage reaches approximately 50 - 70V, the transistor opens like an avalanche, and the capacitor is discharged through transistor VT3 to the base of transistor VT2 and winding III of transformer T2, thereby starting the converter.
UPS design and details
The power supply is assembled on a printed circuit board made of single-sided fiberglass.I don’t provide a drawing of the board, since everyone has their own parts in their stash. I’ll limit myself to just a photo of my board:
In my opinion, there is no point in ironing such a board, it is too simple.
As transistors VT1 and VT2, you can use domestic KT812, KT704, KT838, KT839, KT840, that is, with a collector-emitter boundary voltage of at least 300V; of the imported ones, I know only J13007 and J13009, they are used in computer power supplies. The diodes can be replaced with any other powerful pulsed ones or with a Schottky barrier; for example, I used imported FR302.
Transformer T1 wound on two folded rings K32×19X7 made of ferrite grade M2000NM, the primary winding is wound evenly throughout the entire ring and is 82 turns of PEV-1 0.56 wire. Before winding, it is necessary to round off the sharp edges of the rings with a diamond file or fine sandpaper and wrap them with a layer of fluoroplastic tape, 0.2 mm thick, and also wrap the primary winding. Winding III is wound with PEV-1 0.56 wire folded in half and is 16+16 turns with a tap from the middle. Winding II is wound with two turns of MGTF 0.05 wire, and is located in a place free from winding III.
Transformer T2 wound on a ring K10×6X5 made of ferrite of the same brand. All windings are wound with MGTF 0.05 wire. Winding I consists of ten turns, and windings II and III are wound simultaneously in two wires and make up six turns.
UPS setup
ATTENTION!!! THE PRIMARY CIRCUIT OF THE PSU IS UNDER MAINS VOLTAGE, SO PRECAUTIONS SHOULD BE FOLLOWED WHEN SETTING UP AND OPERATING.
It is advisable to start the unit for the first time by connecting it through a current-limiting resistor, which is an incandescent lamp with a power of 200 W and a voltage of 220 V. As a rule, a properly assembled power supply does not need adjustment, the only exception is the VT3 transistor. You can check the relaxer by connecting the emitter of the transistor to the negative pole. After turning on the unit, sawtooth pulses with a frequency of about 5 Hz should be observed on the transistor collector.
The sound quality depends almost as much on the parameters of the power source as on the amplifier itself, and you should not be negligent in its manufacture. There are more than enough descriptions of calculation methods for standard transformers. Therefore, here is a description of a switching power supply that can be used not only with amplifiers based on the TDA7293 (TDA7294), but also with any other 3H power amplifier.
The basis of this power supply unit (PSU) is a half-bridge driver with an internal oscillator IR2153 (IR2155), designed to control MOSFET and IGBT technology transistors in switching power supplies. The functional diagram of the microcircuits is shown in Figure 1, the dependence of the output frequency on the ratings of the RC-drive chain in Figure 2. The microcircuit provides a pause between the pulses of the “upper” and “lower” switches for 10% of the pulse duration, which allows you not to worry about “through” currents in the power part of the converter.
Rice. 1
Rice. 2
The practical implementation of the power supply is shown in Figure 3. Using this circuit, you can make a power supply with a power from 100 to 500 W, you only need to proportionally increase the capacitance of the primary power filter capacitor C2 and use the corresponding power transformer TV2.
Rice. 1
The capacitance of capacitor C2 is selected at the rate of 1...1.5 µF per 1 W of output power, for example, when manufacturing a 150 W power supply, a capacitor of 150...220 µF should be used. The VD primary power supply diode bridge can be used in accordance with the installed primary power supply filter capacitor; with capacitances up to 330 µF, 4...6 A diode bridges can be used, for example RS407 or RS607. With a capacitor capacity of 470... 680 μF, more powerful diode bridges are needed, for example RS807, RS1007.
We can talk about the manufacture of a transformer for a long time, but not everyone needs to delve into the deep theory of calculations for too long. Therefore, calculations according to Eranosyan’s book for the most popular standard sizes of ferrite rings M2000NM1 are simply summarized in Table 1.
As can be seen from the table, the overall power of a transformer depends not only on the dimensions of the core, but also on the conversion frequency. It is not very logical to make a transformer for frequencies below 40 kHz - harmonics can create insurmountable interference in the audio range. The manufacture of transformers for frequencies above 100 kHz is no longer permissible due to self-heating of the M2000NM1 ferrite by eddy currents. The table shows data on the primary windings, from which the turns/volt ratios can be easily calculated, and then it will not be difficult to calculate how many turns are needed for a particular output voltage. It should be noted that the voltage supplied to the primary winding is 155 V - the mains voltage of 220 V after the rectifier and smoothing filter will be 310 V DC, the circuit is semi-bridged, therefore half of this value will be applied to the primary winding. It should also be remembered that the shape of the output voltage will be rectangular, therefore, after the rectifier and smoothing filter, the voltage value will not differ significantly from the calculated one.
The diameters of the required wires are calculated from a ratio of 5 A per 1 sq mm of wire cross-section. Moreover, it is better to use several wires of smaller diameter than one, thicker wire. This requirement applies to all voltage converters with a conversion frequency above 10 kHz, since the skin effect - losses inside the conductor - is already beginning to affect, since at high frequencies the current no longer flows across the entire cross-section, but along the surface of the conductor, and the higher the frequency, the stronger the effect losses in thick conductors. Therefore, it is not recommended to use conductors thicker than 1 mm in converters with conversion frequencies above 30 kHz. You should also pay attention to the phasing of the windings - incorrectly phasing windings can either damage the power switches or reduce the efficiency of the converter. But let’s return to the power supply shown in Figure 3. The minimum power of this power supply is practically unlimited, so you can make a power supply with 50 W or less. The upper power limit is limited by certain features of the element base.
To obtain higher powers, more powerful MOSFET transistors are required, and the more powerful the transistor, the greater the capacitance of its gate. If the gate capacitance of the power transistor is quite high, then a significant current is required to charge and discharge it. The current of the IR2153 control transistors is quite small (200 mA), therefore, this microcircuit cannot control too powerful power transistors at high conversion frequencies.
Based on the above, it becomes clear that the maximum output power of a converter based on IR2153 cannot be more than 500...600 W at a conversion frequency of 50...70 kHz, since the use of more powerful power transistors at these frequencies quite seriously reduces the reliability of the device. The list of recommended transistors for power switches VT1, VT2 with brief characteristics is summarized in Table 2.
Rectifier diodes of secondary power circuits must have the shortest recovery time and at least two times the voltage reserve and three times the current. The latest requirements are justified by the fact that the self-induction voltage surges of a power transformer amount to 20...50% of the output voltage amplitude. For example, with a secondary power supply of 100 V, the amplitude of self-induction pulses can be 120... 150 V and despite the fact that the duration of the pulses is extremely short, it is enough to cause a breakdown in the diodes, when using diodes with a reverse voltage of 150 V. Threefold reserve current is necessary so that the diodes do not fail at the moment of switching on, since the capacitance of the secondary power filter capacitors is quite high, and quite a small current will be required to charge them. The most suitable diodes VD4-VD11 are summarized in Table 3.
The capacity of the secondary power filters (C11, C12) should not be increased too much, since the conversion is carried out at fairly high frequencies. To reduce ripple, it is much more important to use large capacitance in the primary power circuits and correctly calculate the power of the power transformer. In the secondary circuits, capacitors of 1000 μF per arm are quite sufficient for amplifiers up to 100 W (the power supply capacitors installed on the UMZCH boards themselves must be at least 470 μF) and 4700 μF for a 500 W amplifier. The circuit diagram shows a version of the secondary power supply rectifiers, made on Schottky diodes, and a printed circuit board is installed under them (Figure 4). Diodes VD12, VD13 are used as a rectifier for the forced cooling fan of heat sinks; diodes VD14-VD17 are used as a rectifier for low-voltage power supply (pre-amplifiers, active tone controls, etc.). The same figure shows a drawing of the location of parts and a connection diagram. The converter has overload protection made on the current transformer TV1, consisting of a K20x12x6 ring of M2000 ferrite and containing 3 turns of the primary winding (the cross-section is the same as the primary winding of the power transformer and 3 turns of the secondary winding, wound with a double wire with a diameter of 0.2.. .0.3 mm. If there is an overload, the voltage on the secondary winding of transformer TV1 will become sufficient to open the thyristor VS1 and it will open, closing the power supply to the IR2153 microcircuit, thereby stopping its operation. The protection threshold is adjusted by resistor R8. Adjustment is made without load, starting with maximum sensitivity and achieving stable startup of the converter. The principle of adjustment is based on the fact that at the moment of starting the converter it is loaded to the maximum, since it is necessary to charge the capacitance of the secondary power filters and the load on the power part of the converter is maximum.
About the remaining details: capacitor C5 - film capacitor 0.33... 1 µF 400V; capacitors C9, C10 - film capacitors 0.47...2.2 µF at least 250V; inductances L1...L3 are made on K20x12x6 M2000 ferrite rings and are wound with 0.8...1.0 mm wire until they are filled turn to turn in one layer; C14, C15 - film 0.33...2.2 µF for a voltage of at least 100 V with an output voltage of up to 80 V; capacitors C1, C4, C6, C8 can be ceramic, type K10-73 or K10-17; C7 can also be ceramic, but film, such as K73-17, is better.
